Linear Motor System and Compressor

ABSTRACT

Controllability of a linear motor or a compressor is improved in a linear motor system that includes: an armature having magnetic poles and winding wires; a mover having a permanent magnet; and a power conversion unit that outputs AC power to the winding wires, in which the mover and the armature are relatively movable, and the mover or the armature is connected to an elastic body. The linear motor system further includes: a position detection unit that detects and outputs the position of the mover with respect to the armature, a position estimation, or a current detection unit that outputs the value of current flowing through the winding wires; and a control unit that controls the output of the power conversion unit on the basis of the output of the position detection unit, the output of the position estimation unit, or the output of the current detection unit.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Divisional of U.S. application Ser. No.16/304,600, filed Nov. 26, 2018, which is a 371 of InternationalApplication No. PCT/JP2017/014756, filed Apr. 11, 2017, which claimspriority from Japanese Patent Application No. 2016-105754, filed May 27,2016, the disclosures of which are expressly incorporated by referenceherein.

TECHNICAL FIELD

The present invention relates to linear motor systems and compressors.

BACKGROUND ART

Patent Document 1 discloses a configuration in which a drive voltage Vhaving an angular frequency ω to a linear motor, a phase difference θbetween a counter electromotive force E and the drive voltage Vgenerated in the linear motor is detected, and the angular frequency ωis controlled to make a phase difference ϕ between a drive current I andE practically zero, where ϕ is given by ϕ=tan¹[V sin θ/(V cos θ−E)]−tan¹(ωL/R) using V, ω, the inductance L and the resistance R of the magnetcoil of the linear motor, E, and θ (claim 1, claim 2, and the like ofPLT 1).

CITATION LIST Patent Document

-   Patent Document 1: Japanese Patent Application Laid-Open No.    2002-44977

SUMMARY OF THE INVENTION Technical Problem

In Patent Document 1, the phase difference ϕ is used as an evaluationfunction for controlling ω, and ϕ is dependent on the inductance of thecoil of the linear motor and the like. However, the inductance is avariable dependent on twice the value of the phase of the mover of thelinear motor (2θ). As a result, the evaluation function is a functionincluding θ and 2θ, therefore it is not easy to control w with highaccuracy.

Solution to Problem

A first aspect of the present invention that is achieved with theabove-described problem in mind is a linear motor system that includes:an armature having magnetic poles and winding wires; a mover having apermanent magnet; and a power conversion unit that outputs AC power tothe winding wires, in which the mover and the armature are relativelymovable, and the mover or the armature is connected to an elastic body.The linear motor system further includes: a position detection unit thatdetects and outputs the position of the mover with respect to thearmature, a position estimation unit that estimates and outputs theposition of the mover with respect to the armature, or a currentdetection unit that outputs the value of current flowing through thewinding wires; and a control unit that controls the output of the powerconversion unit on the basis of the output of the position detectionunit, the output of the position estimation unit, or the output of thecurrent detection unit. In the case where a signal having a frequencysubstantially the same as the frequency of the AC power is applied tothe output of the position detection unit, to the output of the positionestimation unit, or to the output of the current detection unit, thecontrol unit changes the frequency of the AC power, and in the casewhere a signal having a frequency substantially larger than thefrequency of the AC power is applied to the output of the positiondetection unit, to the output of the position estimation unit, or to theoutput of the current detection unit, the control unit keeps thefrequency of the AC power substantially the same.

A second aspect of the present invention that is achieved with theabove-described problem in mind is a compressor that includes: anarmature having magnetic poles and winding wires; a mover having apermanent magnet; and a power conversion unit that outputs AC power tothe winding wires, in which the mover and the armature are relativelymovable, the mover or the armature is connected to an elastic body and apiston, and the piston compresses a fluid as a load. The compressorfurther includes: a position detection unit that detects and outputs theposition of the mover or the position of the piston with respect to thearmature, a position estimation unit that estimates and outputs theposition of the mover or the position of the piston with respect to thearmature; and a control unit that controls the output of the powerconversion unit on the basis of the output of the position detectionunit, or the output of the position estimation unit, in which thefrequency of the AC power is increased or decreased in the samecorrelation with the increase or decrease of the load.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a diagram showing a configuration example of a linear motorsystem.

FIG. 2 is a perspective view of a constitutional example of an armature.

FIG. 3 is a schematic diagram showing the cross-sectional view of amagnetic pole and the flows of magnetic fluxes.

FIGS. 4A and B show explanatory diagrams of polarities generated atmagnetic pole teeth.

FIG. 5 is the explanatory diagram of an external mechanism connected toa mover.

FIG. 6 is an explanatory diagram of a relation between a drive frequencyand a stroke.

FIGS. 7A and 7B show explanatory diagrams of a phase relation between anapplied voltage and a current.

FIG. 8 is a vector diagram showing the applied voltage and the current.

FIG. 9 is an explanatory diagram showing a configuration example of aphase difference detector.

FIG. 10 is an explanatory diagram showing a relation between the drivefrequencies and the outputs of a phase difference detector.

FIG. 11 is an explanatory diagram showing a second configuration exampleof a phase difference detector.

FIG. 12 is an explanatory diagram showing a configuration example of adrive frequency adjuster 131.

FIG. 13 is an explanatory diagram showing a configuration example of avoltage instruction value creator 103.

FIG. 14 is a diagram showing a configuration example of a powerconversion circuit.

FIG. 15 is an explanatory diagram showing a configuration example usedin verification.

FIG. 16 is an explanatory diagram showing an example of a superimposedsignal.

FIG. 17 is an explanatory diagram showing a configuration example of alinear motor system according to an embodiment 2.

FIG. 18 is an explanatory diagram showing a configuration example of aphase difference detector according to an embodiment 2.

FIG. 19 is an explanatory diagram showing a relation between a drivefrequency and the output of a phase difference detector according to theembodiment 2.

FIG. 20 is an explanatory diagram showing a configuration example usedin verification according to the embodiment 2.

FIG. 21 is an explanatory diagram showing an example of a superimposedsignal according to the embodiment 2.

FIG. 22 is an example of a vertical cross-sectional view of a hermeticcompressor.

FIG. 23 is an explanatory diagram showing a configuration example of alinear motor system according to an embodiment 3.

FIG. 24 is an explanatory diagram showing a configuration example ofposition estimation means.

FIG. 25 is an operation explanatory diagram used in verificationaccording to the embodiment 3.

DESCRIPTION OF EMBODIMENTS

Hereinafter, examples according to the present invention will beexplained in detail with reference to the accompanying drawings. In theexplanations of the examples, the same components are given the samereference signs, and duplicated explanations about them will be omitted.

It is not necessarily required that various kinds of componentsaccording to the present invention are provided independently of oneanother, and it is allowable that plural components are formed as onemember, one component is composed of plural members, a certain componentis part of another component, part of a certain component and part ofanother component are duplicated, and so on.

Embodiment 1

In this embodiment, although terms “the forth direction” and “the backdirection” that are orthogonal to each other, “the right direction” and“the left direction” that are orthogonal to each other, “the updirection” and “the down direction” that are orthogonal to each otherare used for convenience of explanation, the direction of gravitationalforce is not always equal to the down direction, and it can be setparallel with the forth direction, the back direction, the rightdirection, the left direction, the up direction, the down direction, orother directions.

<Linear Motor Drive Device 101>

FIG. 1 is a schematic diagram of a linear motor system 100. The linearmotor system 100 includes a linear motor drive device 101 and a linearmotor 104. As described later, the linear motor 104 includes an armature9 and a mover 6 that are relatively movable.

The linear motor drive device 101 includes position detection means 106,a control unit 102, and a power conversion circuit 105.

The position detection means 106 detects the relative position of themover 6 (the mover position) to the armature 9.

The control unit 102 outputs an output voltage instruction value to thepower conversion circuit 105, or a drive signal (a pulse signal) fordriving the power conversion circuit 105 according to the detectionresults of the position detection means 106.

Although the detail of the power conversion circuit 105 will bedescribed later, the power conversion circuit 105 is an example of apower conversion unit that converts and outputs the voltage of a DCvoltage source 120. Here, a DC current source can be used instead of theDC voltage source 120.

<Linear Motor 104>

FIG. 2 is a perspective view of the linear motor 104. The armature 9includes two magnetic poles 7 facing each other in both up and downdirections with an air gap therebetween and winding wires 8 windingaround the magnetic poles 7. The mover 6 is disposed in this air gap.The magnetic poles 7 have magnetic pole teeth 70 as their edge surfacesfacing the mover 6.

The armature 9 can give force in the forth or back direction (referredto as thrust hereinafter) to the mover 6. For example, as describedlater, the thrust can be controlled so that the mover 6 can move in theback-and forth direction.

The mover 6 includes two plate permanent magnets 2 (2 a and 2 b) thatare magnetized in up-and-down direction and arranged in the backdirection and the forth direction respectively. In FIG. 2, the permanentmagnets 2 a and 2 b are shown, but the mover 6 is not shown. As themover 6, a plate shaped mover obtained by fixing plate permanent magnets2 can be adopted, for example. The mover 6 is relatively movable in theback-and-forth direction to the armature 9. Hereinafter, the relativevelocity of the mover 6 to the armature is referred to as the velocityof the mover 6.

The control unit 102 can output the drive signal, for example, so thatthe mover 6 can move in the back-and-forth direction within the rangewhere the permanent magnets 2 a and 2 b face the armature 9.

FIG. 3 is a cross-sectional view taken along the line A-A′ of FIG. 2.The arrow lines shown in FIG. 3 show examples of magnetic flux linesthat are generated when a current is flown through the two winding wires8. Because the directions of the flows of the magnetic fluxes can bereversed by the current flowing through the winding wires 8, thedirections of the flows of the magnetic fluxes are not necessarily thedirections shown in FIG. 3. These magnetic flux lines magnetize themagnetic pole teeth 70.

[Thrust Given to Mover 6]

FIG. 4 is a diagram explaining thrust that is exerted on the mover 6 dueto the magnetization of the magnetic pole teeth 70. The polarities ofthe magnetic pole teeth 70 generated by the current flowing through thewinding wires 8 are represented by signs “N” and signs “S” marked downin the vicinities of the magnetic pole teeth 70. FIG. 4(a) shows anexample in which force is exerted on the mover 6 in the forth directionbecause the upper magnetic pole tooth 70 a is magnetized to become “S”and the lower magnetic pole tooth 70 b is magnetized to become “N”. FIG.4(b) shows an example in which force is exerted on the mover 6 in theback direction because the upper magnetic pole tooth 70 a is magnetizedto become “N” and the lower magnetic pole tooth 70 b is magnetized tobecome “S”.

As described above, magnetic circuits including the two magnetic poles 7are provided with magnetic fluxes by applying a voltage or a current tothe winding wires 8, and the two magnetic pole 70 teeth (a pair ofmagnetic pole teeth) facing each other can be magnetized. By applying anAC voltage or an AC current such as a sine wave voltage or current or arectangular wave (a square wave) voltage or current as a voltage or acurrent, thrust that makes the mover 6 move in the back-and-forthdirection can be given to the mover 6. In such a way, the motion of themover 6 can be controlled.

In addition, the thrust given to the mover 6 can be changed by changingthe amplitude of the applied AC voltage or the amplitude of the ACcurrent. Furthermore, the displacement of the mover 6 can be changed ina desired manner by appropriately changing the thrust given to the mover6 using a well-known method. Here, in the case where the mover 6performs back-and-forth movements (for example, these movements aregenerated when the magnetization of the magnetic pole teeth 70 shown inFIG. 4(a) and FIG. 4(b) is sequentially repeated), the change amount ofthe displacement of the mover 6 that changes in an AC waveform manner isreferred to as a stroke.

Because the magnetic pole teeth 70 are magnetic materials, the magneticpole teeth have magnetic sucking forces that draw the permanent magnets2. In this example, because the two magnetic pole teeth 70 are disposedso as to face each other with the mover 6 therebetween, the resultantforce of the magnetic sucking forces exerted on the mover 6 can bereduced.

[Mechanism Outside of Mover 6]

FIG. 5 is an explanation diagram of a mechanism in which an externalmechanism including a resonance spring 23, which is a kind of coilspring, is connected to the mover 6, and the mover 6 is pulled back dueto the spring force of the resonance spring 23. The one end of theresonance spring 23 is connected to the mover 6 via an intermediatesection 24 therebetween, and the other end is fixed to a base section25. In addition, a side section 26 is installed in such a way that theside section 26 extends approximately parallel to the extendingdirection of the resonance spring 23 in order to guide or support theresonance spring 23.

In the case of making the linear motor 104 move in the back-and-forthdirection, acceleration and deceleration are alternately repeated everytime the movement direction of the mover 6 is changed. In the case ofthe deceleration, although the velocity energy of the mover 6 isconverted into electric energy (a regenerative operation), loss isgenerated due to the resistances of wires connected to the linear motor104. On the other hand, as shown in FIG. 5, if the resonance spring 23(the assistant spring) is attached to the mover 6, and the mover 6 ismoved back-and-forth at a mechanical resonance frequency determined bythe mass and spring constant of the mover 6, the velocity energy of themover 6 can be effectively utilized, so that a highly efficient linearmotor drive system can be configured. A well-known elastic body can beused instead of the resonance spring 23. Although, with such aconfiguration of the linear motor drive system, the drive system isconfigured as a mover 6 (a field magneton 6) mobile type drive system inwhich the mover 6 (the field magneton 6) moves with respect to theground, the drive system can be configured as an armature mobile typedrive system in which the armature 9 moves with respect to the groundusing the armature 9 connected to the elastic body instead of the mover6.

FIG. 6 is a diagram showing a relation between the frequency of the ACvoltage (in the vertical axis) and the stroke of the mover 6 (in thehorizontal axis). The amplitude of the AC voltage is constant at eachfrequency. As is clear from FIG. 6, the stroke of the mover 6drastically becomes large in the vicinity of the resonance frequency,and it becomes smaller as the frequency of the AC voltage gets fartheraway from the resonance frequency. Although the resonance frequency isgiven by the square root of the value obtained by dividing the springconstant k of the resonance spring 23 by the mass m of the mover 6, thisvalue may be an approximate value depending on a type of the linearmotor 104.

In such a way, by applying an AC voltage having the resonance frequencyor a frequency in the vicinity of the resonance frequency, it becomespossible to vibrate the mover 6 with a large stroke (with large energy).In other words, it is important to detect or estimate the resonancefrequency of the mover 6. It is preferable to this detection orestimation is accurately executed in order to accurately control thedrive frequency in accordance with the variations of the spring constantk and the mass m of the resonance spring.

[Phase Relation at Driving Time]

FIG. 7(a) is a diagram showing a relation between the position andvelocity of the mover 6 and FIG. 7(b) is a diagram showing a relationbetween the applied voltage waveform and the current flowing through thelinear motor 104 along with the passage of time when the linear motor104 is driven. FIG. 8 is a diagram showing the AC waveforms shown inFIG. 7 as vectors. As is clear from FIG. 8, the phases of the velocityof the mover 6, the applied voltage, and the motor current areapproximately identical. In the case where the resonance spring 23 isadded to the mover 6, and the mover 6 is moved back-and-forth at amechanical resonance frequency determined by the mass and springconstant of the mover 6, it is known that the phase of the position ofthe mover 6 is different from the phase of the applied voltage V to thewinding wires 8, the phase of the motor current Im, and the phase of thevelocity of the mover 6 by 90 degrees respectively. In other words, ifany of the above-mentioned relations is established, it can be estimatedthat the mover 6 is driven at the resonance frequency.

For example, in the case where the mass of the mover 6 departs from theexpected value due to production variations (anomalies), or in the casewhere the mass connected to the resonance spring 23 varies due to a loadelement added to the mover 6, the resonance frequency changes. In orderto obtain the maximum stroke even in such a case, it is preferable tohighly accurately detect or estimate the resonance frequency thatchanges in accordance with some conditions. Hereinafter, the detectionmethod and the estimation method of the resonance frequency will beexplained.

<Control Unit 102>

The control unit 102 and the like will be explained with reference toFIG. 1 and the like. A position detection value xm, which is detected bythe position detection means 106, is input into the control unit 102.The position detection value xm includes information about the phase ofthe mover 6. The input position detection value xm along with a phaseinstruction value θ* generated by the control unit 102 is input into aphase difference detector 130, and a phase difference estimation valuedltθ{circumflex over ( )} is output. A residual error between the phasedifference instruction value dltθ*, which is a target value, and thephase difference estimation value dltθ{circumflex over ( )} is inputinto a drive frequency adjuster 131. The drive frequency adjuster 131outputs a frequency instruction value ω*. The applied voltage V based onthe frequency instruction value ω* is output to the linear motor 104. Asthe position detection means 106, a calculation unit that estimates therelative position using a position sensor that detects the relativeposition of the mover 6 to the armature 9, the applied voltage V, andthe motor current Im, or the like can be adopted. Hereinafter, thecontrol unit 102 and the like will be explained in detail.

The control unit 102 includes: the phase difference detector 130; thedrive frequency adjuster 131 that adjusts a drive frequency instructionvalue ω* on the basis of the output dltθ{circumflex over ( )} of thephase difference detector 130 and the phase difference instruction valuedltθ* so that the phase difference estimation value dltθ{circumflex over( )} follows the phase difference instruction value dltθ*; an integrator140 that integrates the drive frequency instruction value ω* and createsthe phase instruction value θ*; a voltage instruction value creator 103that outputs a voltage instruction value V* on the basis of the phaseinstruction value θ* and the stroke instruction value I* of the mover 6;and a PWM signal creator 133 that outputs a drive signal for driving thepower conversion circuit 105, which outputs a voltage, by comparing thevoltage instruction value V* and a triangular wave carrier signal. Here,the power conversion circuit 105 can be a current output-type powerconversion circuit. In this case, a current instruction value creatorhas only to be installed instead of the voltage instruction valuecreator 103.

<Reference Phase Creator>

In the case where a position sensor is used for estimating the relativeposition xm of the mover 6 to the armature 9, the output of the positionsensor can be used, and therefore a well-known position sensor has onlyto be adopted appropriately. In this example, although means forestimating the resonance frequency using a difference between the phaseof the position of the mover 6 and the phase of the applied voltage V tothe winding wires 8 or the phase of the motor current Im will beexplained, let's first explain a reference phase (the phase of the mover6) that acts as a reference for other phases.

The reference phase (the phase instruction value θ*) of this example isobtained by integrating the drive frequency instruction value ω*, whichis the output of the drive frequency adjuster 131 shown in FIG. 1, usingthe integrator 140 that acts as a reference phase creator. In otherwords, the reference phase is the phase θ* of a wave having the drivefrequency instruction value ω* corresponding to the target frequency ofthe applied voltage V(θ*) at each time. As described above, although theintegral of the drive frequency instruction value ω* of the drivefrequency adjuster 131 is used as the reference phase θ*, it is alsoconceivable that the integrated value is fixed to, for example, themechanical resonance frequency of a vibration body including the mover6.

In the case where the reference phase θ* is used as the phase of theapplied voltage V, the reference phase θ* can also be used in thedetection or estimation of the position of the mover 6. The referencephase θ* can be set to a saw-tooth wave the value range of which is, forexample, [−η, η], [0, 2η], or wider with time, or can be set to alinearly increasing wave with time during a time period where the drivefrequency instruction value ω* is constant. As described later, if thedrive frequency instruction value ω* changes, the shape of the saw-toothwave or the shape of the linearly increasing wave vary (the slopes ofthese waves change).

It goes without saying that the position detection value xm detected bythe position detection means 106 can be used for obtaining the referencephase θ*. In the case of using the position detection value xm, thetotal transfer length of the displacement of one period of theback-and-forth movement of the mover 6 is set to 360°, for example, andthe reference phase θ* can be calculated using the ratio of the position(=displacement) of the mover 6 from the reference position (for example,the intermediate point of the back-and-forth movement, the maximumposition or the minimum position of the back-and-forth movement) to alength corresponding to the total transfer length.

<Phase Difference Detector 130>

In the case where the mover 6 moves in the back-and-forth direction, theposition xm of the mover 6 is represented by a periodic function.Because a periodic function can be represented by a Fourier series, theposition xm of the mover 6 can be given by the following expressionusing a Fourier transform expression.

$\begin{matrix}\lbrack {{Expression}\mspace{14mu} 1} \rbrack & \; \\{\mspace{155mu} {x_{m} = {x_{0} + {\sum\limits_{n = 1}^{\infty}\{ {{a_{n} \cdot {\cos ( {n\; \omega_{0}t} )}} + {b_{n} \cdot {\sin ( {n\; \omega_{0}t} )}}} \}}}}} & (1)\end{matrix}$

Here, x0 is a DC offset value, and an and bn are nth-order Fouriercoefficients and an and bn are given by the following expressions.

$\begin{matrix}\lbrack {{Expression}\mspace{14mu} 2} \rbrack & \; \\{\mspace{225mu} {a_{n} = {\frac{2}{T_{0}} \cdot {\int_{- \frac{\pi}{2}}^{\frac{\pi}{2}}{{x_{m} \cdot {\cos ( {n\; \omega_{0}t} )}}{dt}}}}}} & (2) \\\lbrack {{Expression}\mspace{14mu} 3} \rbrack & \; \\{\mspace{220mu} {b_{n} = {\frac{2}{T_{0}} \cdot {\int_{- \frac{\pi}{2}}^{\frac{\pi}{2}}{{x_{m} \cdot {\sin ( {n\; \omega_{0}t} )}}{dt}}}}}} & (3)\end{matrix}$

Here, T0 is the period of the fundamental wave (the period during whichthe mover 6 performs the back-and-forth movement once); in other words,T0 is the reciprocal of a primary frequency (a drive frequency).

In the case of controlling the mover 6 to be driven at the resonancefrequency, the higher order components are not important; in otherwords, only the primary component, that is, only the drive frequencycomponent had to be noticed. In particular, the phase θ of the primaryfrequency component (the drive frequency component) of the position xmof the mover 6 is important. The phase of the position xm of the mover 6relative to the applied voltage V having a sine wave pattern can begiven by the following expression using arctangent of the first orderFourier coefficients. As described above, it is preferable that thetarget value dltθ* of the phase difference dltθ should be set andcontrolled so that a difference between the phase of the applied voltageV (the reference phase θ*) and the phase of the position xm of the moverbecomes 90°.

$\begin{matrix}\lbrack {{Expression}\mspace{14mu} 4} \rbrack & \; \\{\mspace{140mu} {\theta_{pos}^{\hat{}} = {{\tan^{- 1}( \frac{b_{1}}{a_{1}} )} = {\tan^{- 1}( \frac{\int_{{- 2}\; \pi}^{0}{{x_{m} \cdot \sin}\; \omega_{0}t\; {dt}}}{\int_{{- 2}\; \pi}^{0}{{x_{m} \cdot \cos}\; \omega_{0}t\; {dt}}} )}}}} & (4)\end{matrix}$

In Expression (4), the integration range runs from −2π to 0, and this isbecause only information in the past can be obtained in the case wherethe phase difference detector 130 is materialized by semiconductorintegrated circuits and the like such as microcomputers and DSPs.

FIG. 9 is an explanatory diagram showing a block diagram for bringingExpression (4) into shape. The phase instruction value θ* is input intoa sine calculator 81 (which outputs the sine value of an input value),and a cosine calculator 82 (which outputs the cosine value of an inputvalue) respectively, so that the sine value and the cosine value for thephase instruction value θ* are obtained. The sine value multiplied bythe position xm of the mover 6 and the cosine value multiplied by theposition xm of the mover 6 are output from multipliers 92 respectively.The outputs are respectively integrated by integrators 94 a and 94 b, sothat the first order Fourier coefficient of sine components and thefirst order Fourier coefficient of cosine components are obtained. Inother words, frequency components having frequencies higher than thedrive frequency ω in the Fourier expansion can be eliminated, so thatthe robustness against noises having higher order frequencies can beobtained.

The outputs of the integrators 94 a and 94 b are input into anarctangent calculator 86. The arctangent calculator 86 outputs anarctangent value on the basis of the input sine component and cosinecomponent. Although the arctangent calculator 86 of this example outputsthe arctangent value of a phase the numerator of which is the output ofthe integrator 94 a and the denominator of which is the output of theintegrator 94 b, the arctangent value of a phase the numerator anddenominator of which are exchanged with each other can also be output.FIG. 10 is a diagram showing a relationship between the frequency of theAC voltage (the horizontal axis) and the output value of the arctangentcalculator 86 (the phase difference dltθ) (the vertical axis). As isclear from FIG. 10, even if the values of the integrators 94 a and 94 bare the same, the output value of the arctangent calculator 86 (thephase difference dltθ) changes depending on a combination of thenumerator and the denominator input into the arctangent calculator 86.In the case of this example where a value obtained by multiplying thesine value by the position xm of the mover 6 is set to the numerator anda value obtained by multiplying the cosine value by the position xm ofthe mover 6 is set to the denominator, if the drive frequency is theresonance frequency, 90° is output from the arctangent calculator 86.The value output from the arctangent calculator 86 is larger than 90° ifthe driving frequency is higher than the resonance frequency, andsmaller than 90° if the driving frequency is lower than the resonancefrequency. Herewith, the phase difference dltθ of the primary frequencycomponent of the input AC signal (the position xm of the mover 6 in thisembodiment) input into the phase difference detector 130 relative to thereference phase θ* can be obtained, which makes it possible to estimatethe resonance frequency. It is preferable to control the phasedifference dltθ so that the phase difference dltθ when the referencephase θ* becomes equal to the fundamental frequency θ is set to thetarget value dltθ*. In other words, it is desirable that the targetvalue dltθ* should be appropriately selected depending on a combinationof the numerator and the denominator input into the arctangentcalculator 86.

Because, when inputs (the reference phase θ* and the position xm of themover) change, the convergence times of the integrators 94 a and 94 bare comparatively large, if, for example, an offset signal is added tothe output of the position detector 106 as a superimposed signal, thefrequency of the AC power is apt to fluctuate for a comparatively longtime until the outputs of the integrators 94 a and 94 b converge. Asimilar thing holds true in the case where a signal having a frequencylower than a frequency that is supposed to be the drive frequencyinstruction value ω* is added as a superimposed signal as well.

Therefore, it is conceivable that imperfect integrators are used insteadof the integrators 94 a and 94 b. An imperfect integrator is a kind of alow-pass filter, and the configuration of the imperfect integrator issimilar to that of a primary delay filter. FIG. 11 shows an example of aphase difference detector in which primary delay filters 141 is usedinstead of the integrators 94 a and 94 b. Using the primary delayfilters 141 and setting appropriate time constants T makes it possibleto shorten the convergence time of the phase difference detection. Inother words, in the case where an offset signal is added to the input oroutput of the position detection means 106 as a superimposed signal usedin the after-mentioned verification method, the frequency of the ACpower can be easily kept approximately constant. In other words,robustness can be secured in a system in which disturbances and the likeeasily are superimposed on the position xm of the mover.

Alternatively, it is conceivable that high-pass filters (not shown) areinstalled on paths that are located before the integrators 94 a and 94 b(or the imperfect integrators 141) and through which the signalsmultiplied by the position xm of the mover flow. The cutoff frequenciesof the high-pass filters can be set smaller than the lower limit valueof the drive frequency co; for example, equal to or smaller than 10 Hzor 5 Hz in the case of a linear motor system such as this embodiment.

In this way, in the case where the phase difference detector 130calculates the phase θ of the position xm of the mover corresponding tothe AC voltage instruction value V* using the arctangent of the ratio ofthe first order Fourier coefficients of the drive frequency components,the phase difference detector 130 has a large sensitivity only to theprimary frequency component of an input AC signal input into itself. Inother words, for example, even if a DC offset or higher order noises aresuperimposed on the position xm of the mover 6, the phase differencedltθ of the primary frequency component of the input AC signal into thephase difference detector 130 relative to the reference phase θ* can becalculated more accurately. In addition, if high-pass filters areinstalled as mentioned above, robustness can be secured againstfrequencies lower than the drive frequency co as well.

Therefore, in the case where a system in which noises are easilysuperimposed, such as a system in which an inductance is largelydependent on the position of the mover or a system located near toanother instrument, is adopted as a method for detecting the position ofthe mover 6, especially effective control can be realized. In this way,the resonance frequency can be detected or estimated with high accuracy,and highly efficient linear motor driving can be realized.

<Drive Frequency Adjuster 131>

FIG. 12 is an explanatory diagram showing the configuration of the drivefrequency adjuster 131. The drive frequency adjuster 131 calculates adifference between the phase difference instruction value dltθ* (forexample, 90°) and the phase difference dltθ obtained in the phasedifference detector 130 at a subtractor 91, performs proportionalcontrol over the above difference by multiplying the above difference bya proportional gain Kp_adtr at a multiplier 92 b, performs integralcontrol over the above difference by multiplying the above difference byan integral gain Ki_adtr at a multiplier 92 c and by integrating theresultant value obtained by the above multiplication at an integrator 94c, and adds a calculation result obtained by the proportional controland a calculation result obtained by the integral control using an adder90, and then the drive frequency adjuster 131 outputs the drivefrequency instruction value ω*.

Here, it is conceivable that the phase difference instruction valuedltθ* is obtained using higher-level control means (not shown in thisexample), and it is also conceivable that the difference instructionvalue dltθ* is set to, for example, 90° in advance as shown in thisembodiment. Furthermore, although the drive frequency adjuster 131according to this embodiment is fabricated in aproportional-and-integral control configuration, other controlconfigurations such as a proportional control configuration and anintegral control configuration can also be applied to the drivefrequency adjuster 131.

[Realization of High-Efficiency Drive]

The operations of the phase difference detector 130 and the drivefrequency adjuster 131 in the case where the linear motor 104 is drivenat the mechanical resonance frequency determined by the mass and springconstant of the mover 6 will be explained.

For example, if the mass of the mover 6 is larger than the designedvalue of the mass, the actual resonance frequency becomes lower than thedesigned value of the resonance frequency. In other words, if theinitial value of the drive frequency is determined using the designedvalue of the mass of the mover 6 (if the initial value of the drivefrequency instruction value ω* is determined using the designed value ofthe mass), the linear motor 104 is driven at a frequency higher than theactual resonance frequency. In this case, the phase difference dltθcalculated by the phase difference detector 130 becomes larger than thephase difference instruction value dltθ*. Therefore, the drive frequencyadjuster 131 performs control to reduce the drive frequency instructionvalue ω*, so that the drive frequency instruction value ω* coincideswith an actual resonance frequency. As a result, the velocity energy ofthe mover 6 can be effectively utilized, and the linear motor 104 canefficiently be driven.

<Voltage Instruction Value Creator 103>

Hereinafter, a configuration, in which the linear motor 104 is driven bya voltage V that is created on the basis of a voltage instruction valueVm* and applied to the linear motor 104, will be explained.

FIG. 13 is an explanatory diagram showing the configuration of a voltageinstruction value creator 103. The voltage instruction value creator 103brings in the phase instruction value θ* and the stroke instructionvalue I*, and outputs the single-phase AC voltage instruction value Vm*.The details will be given below.

In this embodiment, an assumption that the value obtained by multiplyingthe stroke instruction value I* by the sine of the reference phase θ*(sin θ*) is set to the position instruction value xm* of the mover 6 isadopted. First, the phase instruction value θ* is input into the cosinecalculator 82 b (which outputs the cosine of the input value), and thecosine of the phase instruction value θ* (cos θ*) is obtained. Thiscosine value, the stroke instruction value I*, and the frequencyinstruction value ω* are multiplied at a multiplier 92 d. With such aconfiguration, it is possible to obtain the velocity instruction valuevm* of the mover 6 without executing differential calculation. Generallyspeaking, it is possible to make a sine of one of the positioninstruction value xm* and the velocity instruction value vm* and acosine of the other.

In addition, the single-phase AC voltage instruction value Vm* isobtained by multiplying the velocity instruction value vm* of the mover6 by an induced voltage constant Ke* at a multiplier 92 e.

Furthermore, a well-known drive voltage instruction method for driving asynchronous motor can also be applied to the voltage instruction valuecreator 103 instead of the above method.

<PMW Signal Creator 133>

In the PWM signal creator 133, a drive signal corresponding to thevoltage instruction value Vm* is generated using a well-known pulsewidth modulation in which a triangular wave carrier signal and thevoltage instruction value Vm* are compared with each other.

<Power Conversion Circuit 105>

FIG. 14 is an explanatory diagram showing the configuration of the powerconversion circuit 105. The power conversion circuit 105 includes the DCvoltage source 120, a shunt resistor 125, and a full bridge circuit 126.The full bridge circuit 126 outputs a voltage to the linear motor 104 byswitching the DC voltage source 120 in accordance with a drive signalinput from the control unit 102. The full bridge circuit 126 includesfour switching elements 122, and the four switching elements 122 form afirst upper and a first lower arms (referred to as a U-phasehereinafter) having switching elements 122 a and 122 b respectively anda second upper and a second lower arms (referred to as a V-phasehereinafter) having switching elements 122 c and 122 d respectively. Theswitching elements 122 can execute switching operations in accordancewith pulse-shaped gate signals (124 a to 124 d) that a gate drivercircuit 123 outputs on the basis of a voltage instruction valuegenerated by the control unit 102 and a drive signal generated by pulsewidth modulation.

By controlling the conduction states (ON/OFF) of the switching elements122, the DC voltage of the DC voltage source 120 the voltagecorresponding to an AC voltage can be output into winding wires 8. Here,a DC current source can be used instead of the DC voltage source 120. Asthe switching elements 122, semiconductor switching elements such asIGBTs and MOS-FETs can be adopted, for example.

[Wire Connection to Linear Motor 104]

In the power conversion circuit 105, a point between the switchingelement 122 a of the first upper arm and the switching element 122 b ofthe first lower arm is connected to the linear motor 104 and a pointbetween the switching element 122 c of the second upper arm and theswitching element 122 d of the second lower arm is also connected to thelinear motor 104. In FIG. 14, although the winding wires 8 of the upperand lower armatures 9 are connected in parallel with each other, it isalso possible to connect the winding wires 8 in series.

[Current Detection Means 107]

Current detection means 107 such as CTs (current transformers) can beinstalled on the U-phase lower arm and V-phase lower arm. Using theseCTs, a motor current Im flowing through the winding wires of the linearmotor 104 can be detected.

As the current detection means 107, a phase shunt current scheme, inwhich shunt resistors are installed to the lower arms of the powerconversion circuit 105 instead of the CTs, and a current flowing throughthe linear motor 104 is detected from currents flowing through the shuntresisters, can be adopted. It is also conceivable that, instead of or inaddition to the current detection means 107, a single shunt currentdetection scheme, in which a current in the AC side of the powerconversion circuit 105 is detected from a DC current flowing through ashunt resistor 125 installed in the DC side of the power conversioncircuit 105, is adopted. In the single shunt current detection scheme, aphenomenon, in which the current flowing through the shunt resistor 125changes with time depending on the conduction states of the switchingelements 122 included in the power conversion circuit 105, is utilized.

[Verification Method]

In the linear motor drive system according to this embodiment, there aremany cases where the control unit 102 includes semiconductor integratedcircuits (calculation control means) such as microcomputers or DSPs, andthe functions of the control unit 102 are realized by software or thelike. A method for verifying whether the control unit 102 is correctlyconfigured or not will be explained below.

FIG. 15 is a diagram for explaining an example of a circuitconfiguration for verifying the operations of the phase differencedetector 130 and the drive frequency adjuster 131. As shown in FIG. 15,the circuit configuration is made so that a verifier can add asuperimposed signal to the position detection value xm. Although thecircuit shown in FIG. 15 is configured so that the superimposed signalis added to the output of the position detection means 106, it is alsoconceivable that the superimposed signal is added to the input into theposition detection means 106.

In this verification method, in the case where a signal, which iscreated by adding an after-described superimposed signal to the positiondetection value xm output from the position detection means 106, isinput into the control unit 102, this signal is freshly referred to as aposition detection value xm. FIG. 16 is a diagram for explaining thechange of the position detection value xm when the superimposed signalis changed.

Time Period 1 is a time period during which any superimposed signal isnot added to the output of the position detection means 130. In the casewhere the mover 6 performs symmetrical back-and forth movements in TimePeriod 1, the displacement of the mover 6 (the position detection valuexm of the position detection means 106) changes in a sine wave patternassuming that the center position of the back-and-forth movement iszero. Therefore, during Time Period 1 of FIG. 16, a waveform of a sinewave pattern is obtained as the position detection value xm. Assumingthat the phase difference detector 130, which is configured as mentionedabove, is used and the phase difference instruction value dltθ* is 90°,the drive frequency during Time Period 1 approaches the resonancefrequency.

Time Period 2 is a time period during which a DC offset is added to theoutput of the position detection means 130 as a superimposed signal.Because the phase difference detector 130 has a large sensitivity onlyto the primary frequency component of the input AC signal xm because ofusing integrators or imperfect integrators having appropriate timeconstants, the phase difference output by the phase difference detector130 during Time Period 2 is not so much different from the phasedifference during Time Period 1. Therefore, the drive frequency hardlychanges. In this case, it is preferable that the imperfect integratorsshould be used because the convergence time becomes shorter. Inaddition, it is preferable that high-pass filters should be used becausethe convergence time becomes shorter even if signals of lower orderfrequencies are added.

Time Period 3 is a time period during which a signal having the samefrequency component as the drive frequency is added to the output of theposition detection means 130 as a superimposed signal. When the signalhaving the same frequency component is superimposed, the waveform of asignal (the synthesized waveform) input into the control unit 102 has adifferent phase relative to the waveform of the output of the positiondetection means 130. In other words, the control unit 102 judges thatthe phase difference dltθ departs from the instruction value dltθ*, sothat the drive frequency instruction value ω* is changed. Therefore, thedrive frequency for the mover 6 changes, and the frequency of thedetection value xm of the position detection means 106 changes. FIG. 16shows that the drive frequency becomes lower.

Time Period 4 is a time period during which a signal having a higherorder frequency than the drive frequency is added to the output of theposition detection means 130 as a superimposed signal. Because the phasedifference detector 130 includes the integrators 94 or the imperfectintegrators 141, the phase difference detector 130 has a largesensitivity only to the primary frequency component of the input ACsignal. Therefore, the phase difference output from the phase differencedetector 130 during Time Period 4 is almost the same as the phasedifferences during Time Period 1 and Time Period 2. Therefore, owing tothe change of the drive frequency executed by the drive frequencyadjuster 131, the drive frequency during Time Period 4 changes to thesame frequency as the frequencies during Time Period 1 and Time Period2. In the case where a voltage V having a frequency approximately equalto the resonance frequency is applied to the mover 6, a signal having afrequency substantially higher than the resonance frequency, forexample, by 10% or 20% can be considered to be a signal having a higherorder frequency.

In this way, by checking a relation between the frequency of the steadystate of the position detection value xm obtained during Time Period 1and the frequencies of the position detection value xm obtained duringpart or all of Time Period 2 to Time Period 4, it becomes possible toverify whether or not the control unit 102 is performing controlaccording to this example in which the drive frequency instruction valueω* is controlled using the first order Fourier coefficients. As is clearfrom the above description, by performing control using the Fourierexpansion of the position xm of the mover and the integration value ofthe product of the sine of the reference phase θ* and the cosine of thereference phase θ* as shown in this example, frequency characteristicsshown especially during Time Period 1, Time Period 3, and Time Period 4can be brought about. Furthermore, by installing imperfect integratorsand/or high-pass filters in the phase difference detector 130, acharacteristic in response to an offset signal or a low frequency signalcan be effectively brought about as shown in Time Period 2.

Here, even if a signal having the same frequency component as the drivefrequency and a signal having a higher order frequency component thanthe drive frequency are added just after Time Period 1 as superimposedsignals, a waveform that is similar to the waveforms obtained duringTime Period 3 and Time Period 4 can be obtained.

[In the Case of Detent Being Large]

In the case where the detent force of the linear motor 104(corresponding to the reluctance torque of a rotary motor) is large,there is a possibility that the phase of the applied voltage V and thephase of the motor current Im that provide the maximum efficiency arenot in phase with each other. In such a case, if the phase instructionvalue θ* is changed to a value determined in consideration of theinfluence of the detent force, the above-described effect can beobtained.

In addition, if a load is large, magnetic saturation occurs at thearmature 9, and the detent force of the linear motor 104 also changes.Therefore, by changing the phase instruction value θ* in accordance withthe magnitude of the load, the velocity energy of the mover 6 caneffectively be used even in the wide range of load condition, so thatthe linear motor 104 can be driven highly efficiently. The appropriatevalue of the phase instruction value θ* is dependent on the detent forcethat changes in accordance with the disposition of magnetic materials inthe structures of the armature 9 and the mover 6 and the like.Consequently, it is preferable that the phase instruction valueadjuster, which adjusts the phase instruction value θ* by detecting orestimating part or all of the magnetic saturation, the motor current Im,the load, and the detent force, should be provided.

Embodiment 2

The configuration of this example can be made just like that of theembodiment 1 except for the following point. In this embodiment, aresonance frequency is estimated using a motor current Im instead ofdetecting or estimating the position xm of a mover 6.

<Linear Motor Drive Device 201>

FIG. 17 is a schematic diagram showing a linear motor system 200. Thelinear motor system 200 includes a linear motor drive device 201 and alinear motor 104.

The linear motor drive device 201 includes: position estimation means208; current detection means 207; a control unit 202; a phase differencedetector 230; and a power conversion circuit 105.

<Phase Difference Detector 230>

FIG. 18 is an explanatory diagram showing a configuration embodiment ofa phase difference detector 230 according to this example, and FIG. 19is a diagram showing a relation between the frequency of an AC voltage(the horizontal axis) and the output value of an arctangent calculator86 (the phase difference dltθ{circumflex over ( )}) (the vertical axis).A phase instruction value θ* is input to a sine calculator 81 (thatoutputs the sine of an input value) and to a cosine calculator 82 (thatoutputs the cosine of an input value) respectively, so that the sine ofthe phase instruction value θ* and the cosine of the phase instructionvalue θ* are obtained. A value obtained by multiplying the sine by themotor current Im and a value obtained by multiplying the cosine by themotor current Im are respectively output from multipliers 92. Afterthose outputs are respectively calculated by primary delay filters 141 aand 141 b, the first order Fourier coefficients of the sine and cosineare respectively obtained. In other words, because the frequencycomponents of the Fourier expansion the frequencies of which are higherthan the drive frequency ω can be eliminated, the linear motor system200 can be configured to be robust against high order noises. Reversecalculation is executed on the sign of the output of the sine calculator81. In this embodiment, although a relation between the phase differencedltθ{circumflex over ( )} and the drive frequency corresponding to“numerator: negative value of integrator 94 b, denominator: integrator94 a” is obtained, other relations can also be adopted. In thisembodiment, because the reference phase θ* and the motor current Im areused, the target value dltθ* of the phase difference can be set to, forexample, 0°.

The outputs of the primary delay filters 141 a and 141 b are input intothe arctangent calculator 86. The arctangent calculator 86 output anarctangent value on the basis of the input sine and cosine components.The arctangent calculator 86 of this embodiment outputs an arctangentvalue of a phase having the negative value of the output of the primarydelay filter 141 a as a numerator and the output of the primary delayfilter 141 b as a denominator. It goes without saying that a valueobtained by using the numerator and the denominator replaced with eachother can also be usable as explained in the example 1.

As is clear from FIG. 19, even if the outputs of the primary delayfilters 141 a and 141 b remain intact, the output value (the phasedifference dltθ{circumflex over ( )}) of the arctangent calculator 86changes depending on how to combine the numerator and the denominatorwhich are input into the arctangent calculator 86. In this embodiment,if the drive frequency is the resonance frequency, 0° is output from thearctangent calculator 86. The value output from the arctangentcalculator 86 is larger than 0° if the drive frequency is higher thanthe resonance frequency, and the value output from the arctangentcalculator 86 is smaller than 0° if the drive frequency is lower thanthe resonance frequency. With this, it becomes possible to calculate thephase difference dltθ{circumflex over ( )} of the primary frequencycomponent of the input AC signal (the position xm of the mover 6 in thisembodiment) that t is input into the phase difference detector 230relative to the reference phase θ*.

As shown in this embodiment, control can also be performed with the useof the product of the sine of the reference phase θ* and the motorcurrent Im, and the product of the cosine of the reference phase θ* andthe motor current Im.

[Verification Method]

A method for verifying whether the control unit 202 is correctlyconfigured or not will be explained below.

FIG. 20 is a diagram for explaining an example of a circuitconfiguration for verifying the operations of the phase differencedetector 230 and the drive frequency adjuster 131. As shown in FIG. 20,this circuit is configured in such a way that a verifier can add asuperimposed signal to the motor current Im. Although the circuit shownin FIG. 20 is configured so that the superimposed signal is added to theoutput of the current detection means 207, it is also conceivable thatthe superimposed signal is added to the input of the current detectionmeans 207.

In this verification method, in the case where a signal, which iscreated by adding an after-described superimposed signal to the motorcurrent value Im output from the current detection means 207, is inputinto the control unit 202, this signal is freshly referred to as a motorcurrent Im. FIG. 21 is a diagram for explaining the change of the motorcurrent Im when the superimposed signal is changed. The superimposedsignal can be considered to be a pseudo disturbance.

Time Period 1 is a time period during which any superimposed signal isnot added to the output of the current detection means 207. In the casewhere the mover 6 performs symmetrical back-and forth movements in TimePeriod 1, the motor current Im changes in a sine wave pattern. Assumingthat the phase difference detector 130, which is configured as mentionedabove, is used and the phase difference instruction value dltθ* is 0°,the drive frequency during Time Period 1 approaches the resonancefrequency.

Time Period 2 is a time period during which a DC offset is added to theoutput of the current detection means 207 as a superimposed signal.Because the phase difference detector 230 has a large sensitivity onlyto the primary frequency component of the input motor current Im in thecase of using imperfect integrators having appropriate time constants orhigh-pass filters, the phase difference output by the phase differencedetector 230 during Time Period 2 is not so much different from thephase difference during Time Period 1. Therefore, the drive frequencyhardly changes.

Time Period 3 is a time period during which a signal having the samefrequency component as the drive frequency is added to the output of thecurrent detection means 207 as a superimposed signal. When the signalhaving the same frequency component is superimposed, the waveform of asignal (a synthesized signal) input into the control unit 202 has adifferent phase relative to the waveform of the output of the currentdetection means 207. In other words, the control unit 202 judges thatthe phase difference dltθ{circumflex over ( )} departs from theinstruction value dltθ*, so that the drive frequency instruction valueω* is changed. Therefore, the drive frequency for the mover 6 changes.FIG. 21 shows that the drive frequency becomes lower.

Time Period 4 is a time period during which a signal having a higherorder frequency than the drive frequency is added to the output of thecurrent detection means 207 as a superimposed signal. Because the phasedifference detector 230 includes the integrators 94 or the imperfectintegrators 141, the phase difference detector 230 has a largesensitivity only to the primary frequency component of the input ACsignal. Therefore, the phase difference output from the phase differencedetector 230 during Time Period 4 is almost the same as the phasedifferences during Time Period 1 and Time Period 2. Therefore, owing tothe change of the drive frequency executed by the drive frequencyadjuster 131, the drive frequency during Time Period 4 changes to thesame frequency as the frequencies during Time Period 1 and Time Period2.

In this way, by checking a relation between the frequency of the steadystate of the position detection value xm obtained during Time Period 1and the frequencies of the position detection values xm obtained duringpart or all of Time Period 2 to Time Period 4, it becomes possible toverify whether or not the control unit 202 is performing controlaccording to this embodiment in which the drive frequency instructionvalue ω* is controlled using the first order Fourier coefficients.

Embodiment 3

The configuration of this embodiment can be made just like that of theembodiment 1 or that of the embodiment 2 except for the following point.This embodiment relates to a hermetic compressor 50 as an example of aninstrument on which a linear motor system 300 is mounted. As such aninstrument, an instrument that provides a vibration body (a mover 6)moving back and forth with a load that changes in accordance with thephase θ and drive frequency ω of the vibration body, or the like can beused.

<Hermetic Compressor 50>

FIG. 22 is an example of a vertical cross-sectional view of the hermeticcompressor 50 including a linear motor 104. The hermetic compressor 50is a reciprocating compressor including an airtight container 3 in whicha compressed component 20 and an electromotive component 30 aredisposed. The compressed component 20 and the electromotive component 30are elastically supported by support springs 49 in the airtightcontainer 3. The electromotive component 30 includes the mover 6 and anarmature 9.

The compressed component 20 includes: a cylinder block 1 that forms acylinder 1 a; a cylinder head 16 that is assembled on one end face ofthe cylinder block 1; and a head cover 17 that forms a discharge chamberspace. A working fluid supplied into the inside of the cylinder 1 a iscompressed by the back-and-forth movement of a piston 4, and thecompressed working fluid is sent to a discharge pipe (not shown) whichcommunicates with the outside of the compressor.

The piston 4 is fixed on one end of the mover 6. In this embodiment, theworking fluid is compressed by the back-and-forth movement of the mover6 and the piston 4. The compressed component 20 is disposed on one endof the electromotive component 30. The cylinder block 1 includes a guiderod, which guides the back-and-forth movement of the mover 6, along theback-and-forth direction.

In the case where the linear motor 104 is installed in the airtightcontainer 3, connectors having airtightness, which are referred to ashermetic connectors or hermetic seals, are sometimes used. In order toretain airtightness, it is desirable that the number of the connectorsshould be minimum. Therefore, in the linear motor system 300 accordingto this embodiment, the position of the mover 6 is estimated from avoltage V applied to the linear motor 104 and a motor current Im flowingthrough the linear motor 104, and a resonance frequency is detected orestimated with high accuracy on the basis of the position estimationvalue xm{circumflex over ( )}, so that high-efficiency linear motordrive is provided.

In the case where a resonance spring 23 (not shown in FIG. 22) is addedto the mover 6, and the mover 6 is moved back-and-forth at a mechanicalresonance frequency determined by the mass and spring constant of themover 6, it is necessary to take the influence of the compressedcomponent 20 on the resonance frequency into consideration. In otherwords, because the spring-like operation of the working fluid is addeddue to the pressure of the discharge space, the frequency that will be aresonance frequency changes. More specifically, the fact that thepressure of the cylinder 1 a is high is equivalent to the fact that thespring constant of the resonance spring 23 added to the mover 6 is high,so that the resonance frequency becomes high. Contrarily, if thepressure of the cylinder 1 a is low, the spring constant of theresonance spring 23 added to the mover 6 becomes dominant, so that theresonance frequency becomes near to the mechanical resonance frequencydetermined by the mass and spring constant of the mover 6.

As mentioned above, in the case where the linear motor 104 gives motiveenergy to the compressed component 20, the resonance frequency changesdepending on the condition of the compressed component 20. In order toobtain the maximum stroke even in such a case, it is necessary to detector estimate the resonance frequency that changes depending on thecondition with high accuracy. Therefore, in the linear motor system 300according to this embodiment, the position of the mover 6 is estimatedfrom the voltage applied to the linear motor 104 and the current flowingthrough the linear motor 104, and the resonance frequency is detected orestimated with high accuracy on the basis of the position estimationvalue, so that high-efficiency linear motor drive can be realized.

<Linear Motor Control Device 301>

FIG. 23 is the schematic diagram of the linear motor system 300. Thelinear motor system 300 includes a linear motor control device 301 andthe linear motor 104.

The linear motor control device 301 includes: position estimation means308; current detection means 307; a control unit 302; a phase differencedetector 330; and a power conversion circuit 105.

<Position Estimation Means 308>

The position estimation means 308 estimates the position of the mover 6.For example, the position estimation means 308 calculates the positionestimation value xm{circumflex over ( )} from the following expressionusing a voltage Vm* applied to the linear motor 104 and the current Imflowing through the linear motor 104.

$\begin{matrix}\lbrack {{Expression}\mspace{14mu} 5} \rbrack & \; \\{\mspace{166mu} {x_{m}^{\hat{}} = {\frac{1}{K_{e}^{*}}\{ {{\int{( {V_{m}^{*} - {R_{m}^{*} \cdot I_{m}}} ){dt}}} - {L_{m}^{*} \cdot I_{m}}} \}}}} & (5)\end{matrix}$

In Expression 5, Vm* is a voltage instruction value Vm* to be applied tothe linear motor 104. Here, it is also conceivable that the positionestimation means 308 estimates the position of the piston 4 instead ofthe position of the mover 6.

FIG. 24 is an explanatory diagram showing a block diagram formaterializing Expression 5. Here, the position estimation means 308 canbe realized by adopting a well-known method for estimating the positionof a synchronous motor instead of the above described method.

As described above, the position of the mover 6 is estimated from thevoltage applied to the linear motor 104 and the current flowing throughthe linear motor 104, and the resonance frequency is detected orestimated with high accuracy on the basis of the position estimationvalue, so that high-efficiency linear motor drive can be provided.

[Verification Method]

A method for verifying whether the control unit 302 is correctlyconfigured or not will be explained below.

FIG. 25 is an operation explanatory diagram of the phase differencedetector 330 and a drive frequency adjuster 131, and shows the timechange of the load of the compressed component 20 and the time change ofthe drive frequency of the linear motor 104.

When the linear motor 104 is driven, the initial value of the drivefrequency is set to, for example, a resonance frequency determined themass and spring constant of the mover 6. When the linear motor 104 isdriven, the working fluid is compressed, so that the spring-likemovement of the working fluid is added and the resonance frequencybecomes high. Because the resonance frequency changes in accordance withthe compressed state of the working fluid, the phase θ of the mover 6departs from a reference phase θ*. Therefore, a residual error between aphase difference dltθ and a phase difference instruction value dltθ*changes. In consideration of this residual error, the drive frequencyadjuster 131 changes a drive frequency instruction value ω*, and adjuststhe drive frequency so that the drive frequency approaches the changedresonance frequency (a frequency higher than the initial frequency).

Time Period 1 is a time period during which the load of the compressedcomponent 20 is gradually increased. Because the resonance frequencygradually increases along with the increase of the load, the drivefrequency instruction value ω* gradually increases, and as a result, thedrive frequency ω gradually increases.

Time Period 2 is a time period during which the value of the load of thecompressed component 20 is kept an almost constant value smaller thanthe final value of the load during Time Period 1. Lowering the value ofthe load of the compressed component 20 causes the spring-like operationof the working fluid to be weakened, and as a result, the resonancefrequency becomes lower than the final value of the resonance frequencyduring Time Period 1. Therefore, the drive frequency adjuster 131adjusts the drive frequency to the changed resonance frequency (a lowerfrequency).

Time Period 3 is a time period during which the value of the load of thecompressed component 20 is kept an almost constant value larger than thefinal value of the load during Time Period 1. Heightening the value ofthe load of the compressed component 20 causes the spring-like movementof the working fluid to be strengthened, and as a result, the resonancefrequency becomes higher than the final value of the resonance frequencyduring Time Period 1. Therefore, the drive frequency adjuster 131adjusts the drive frequency to the changed resonance frequency (afrequency higher than the drive frequency during Time Period 1).

As described above, by checking the change of the drive frequency alongwith the change of the load of the compressed component 20, it becomespossible to verify whether the control unit 202 includes a configurationcompliant with this embodiment or not. According to this embodiment, thedrive frequency can be increased or decreased in the same correlation asthe increase or decrease of the load. In other words, control thatfollows a gas spring constant, which increases or decreases inaccordance with the load, can be performed.

Here, as for the load of the compressed component 20, not only thepressure but also the temperature and the discharge amount can bemeasured instead of the load. In other words, the axis representing theload in FIG. 25 can be viewed as the axis representing the pressure,temperature, or the discharge amount. According to this embodiment, thesame advantageous effect as the example 1 can be achieved.

The present invention is not limited to the above embodiments, andvarious modification examples can be included in the present invention.For example, the above embodiments have been described in detail inorder to make the present invention easily understood, and therefore allthe components described so far are not always indispensable for thepresent invention.

Furthermore, it is conceivable that part or the entirety of each of theabove-described configurations, functions, processing units, processingprocedures, and the like is realized by hardware, for example, throughdesigning the part or entirety using integrated circuits. Alternatively,it is also conceivable that the above-described configurations,functions, and the like are realized by software through the operationsof processors in which the processors interpret programs, which realizethe respective functions, and executes the programs.

REFERENCE SIGNS LIST

-   1—Cylinder Block-   1 a—Cylinder-   2—Permanent Magnet-   3—Airtight Container-   4—Piston-   6—Mover-   7—Magnetic Pole-   8—Winding Wire-   9—Armature-   16—Cylinder Head-   17—Head Cover-   20—Compressed Component-   23—Resonance Spring (Assistant Spring)-   30—Electromotive Component-   50—Hermetic Compressor-   100—Linear Motor System-   101—Linear Motor Drive Device-   102—Control Unit-   103—Voltage Instruction Value Creator-   104—Linear Motor-   105—Power Conversion Circuit-   107, 207—Current detection Means-   122—Switching Element-   126—Full Bridge Circuit-   130, 230, 330—Phase Difference Detector-   131—Drive Frequency Adjuster-   133—PWM Signal Creator-   201, 301—Linear Motor Control Device

1. A compressor including an armature having magnetic poles and windingwires; a mover having a permanent magnet; a power conversion unit thatoutputs AC power to the winding wires, in which the mover and thearmature are movable relative to each other, the mover or the armatureis connected to an elastic body and a piston, and the piston compressesa fluid and a load caused by compressing the fluid changes in accordancewith a phase or a drive frequency of the mover or the armature, thecompressor comprising: a position detection unit that detects andoutputs a position of the mover or a position of the piston with respectto the armature, or a position estimation unit that estimates andoutputs the position of the mover or the position of the piston withrespect to the armature; and a control unit that controls an output ofthe power conversion unit based on an output of the position detectionunit or an output of the position estimation unit, wherein the load ismeasured as pressure, temperature or discharge amount of the fluid and afrequency of the AC power is increased or decreased in correlation withthe increase or decrease of the load.
 2. The compressor according toclaim 2, wherein the frequency of the AC power is controlled using anarctangent of a ratio of Fourier coefficients of the position of themover with respect to the armature.